Introduction

Metasurfaces, artificial composite materials composed of sub-wavelength structures, offer advantages such as compact size, low loss, and a strong ability to control electromagnetic wave properties1,2. Metasurfaces utilize the electromagnetic (EM) coupling effect of structural units to effectively regulate the amplitude and phase of EM waves3, enabling functionalities including polarization manipulation, absorption, transmission, and scattering4,5,6,7. Therefore, metasurfaces have been widely applied in areas such as antenna design, wireless communication, radar technology, and electromagnetic invisibility8,9,10,11.

Recent research demonstrates that metasurfaces have significant potential for electromagnetic wave polarization conversion and electromagnetic stealth. Reference12 proposes a reflective polarization conversion metasurface, that utilizes PIN diodes to switch between cross-polarization and linear-to-circular polarization modes. In13, a cross-polarization converter based on a sandwiched structure is presented, capable of performing both transmission and reflection polarization conversion across different operating frequency bands. In reference14, a metasurface based on surface circuit design is proposed, capable of achieving both polarization conversion and retroreflection simultaneously. The radar cross-section (RCS) is a critical parameter for assessing the effectiveness of target stealth capability. We propose a method to reduce RCS using polarization conversion metasurfaces based on the principle of phase cancellation interference through cross-polarization conversion15,16. In reference17, by adopting different arrangement methods, a reduction in RCS was achieved while maintaining the polarization conversion bandwidth. In18, a metasurface based on cross-polarization conversion is proposed, which achieves beam control and reduction of RCS through 8-bit encoding units based on PB phase, playing a crucial role in manipulation of polarized waves. Additionally, in19, a multi-layer metasurface design for ultra-wideband polarization conversion and reduction of RCS is proposed. However, these metasurface designs suffer from issues such as narrow bandwidth, low flexibility, and structural complexity, which limits their widespread application in practice.

To address the challenges posed by the current complex electromagnetic environment, we propose an innovative design that integrates polarization conversion and EM scattering control into a single metasurface. Figure 1 illustrates the metasurface structure and its functional applications. By adjusting the angle α between the metallic resonant rings, the metasurface enables two polarization conversion mechanisms: linear-to-circular polarization conversion and cross-polarization conversion. Furthermore, based on the phase interference cancellation principle of the polarization conversion metasurface, the characteristics of the scattered beams under different coding arrays are investigated, successfully reducing the target RCS by more than 10 dB. Compared with existing studies, this design offers significant advantages, including ultra-wideband performance, high flexibility, and multifunctionality20,21,22.

Fig. 1
figure 1

Conceptual diagram of multifunctional metasurface design (a) α = 105°, achieving linear to circular polarization conversion, (b) α = 230°, achieving cross-polarization conversion, (c) achieving beam control and RCS reduction.

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Structural design and analysis

Structural analysis

The designed metasurface unit consists of external and internal metallic resonant rings, both of which are arranged in an arc shape. By changing α, the angle between the two metallic resonant rings, various polarization conversion functions are realized. In Fig. 2a, the variation of the angle α is described with α1 = 50°, α2 = 105°, α3 = 170°, and α4 = 230°. By adjusting the angle α of the resonant rings over the 0—360° range, this study evaluates the reflection efficiency of cross-polarization and linear-to-circular polarization conversion, using the polarization conversion ratio (PCR) and axial ratio (AR), respectively When the PCR value exceeds 0.9, the incident linearly polarized wave is converted into a linearly polarized wave in a different direction, indicating excellent performance. Similarly, when the AR is less than 3 dB, a linearly polarized wave can be successfully converted into a circularly polarized wave. Specifically, \(PCR = {{\left| {R_{yx} } \right|^{2} }/{\left| {R_{yx} } \right|^{2} + \left| {R_{xx} } \right|^{2} }} = {{\left| {R_{xy} } \right|^{2} } /{\left| {R_{xy} } \right|^{2} + \left| {R_{yy} } \right|^{2} }}\),and \(AR = \left( {{{\left( {r_{xy}^{2} + r_{yy}^{2} + \sqrt a } \right)} /{\left( {r_{xy}^{2} + r_{yy}^{2} - \sqrt a } \right)}}} \right)^{{{1 / 2}}}\)23. In the frequency range of 12–35 GHz, when the angle α is within the range of 120°–360°, the PCR exceeds 0.9 across multiple bands, successfully achieving cross-polarization conversion, as shown by the red regions in Fig. 2b. Additionally, within the range of α = 60°-120°, linear-to-circular polarization conversion is achieved, with an AR of less than 3 dB, demonstrating multiple wideband characteristics, as indicated by the blue regions in Fig. 2c. This paper selected rotation angles of α = 105° and 230° for research, as these angles provide maximum bandwidth for polarization conversion, with a common bandwidth of 14.6–26.8 GHz. In practical applications, any angle within the 0 to 2π range can be selected to achieve specific frequency bands and functions within the bandwidth of 12–35 GHz.

Fig. 2
figure 2

(a) Schematic diagram of rotating the angle α of the metal resonant rings, (b) PCR corresponding to the angle α, (c) AR corresponding to the angle α.

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Design of metasurface unit

The metasurface designs for different polarization conversion proposed in this paper are illustrated in Fig. 3. Figure 3a shows the structure of the linear-to-circular polarization conversion metasurfaces (LTC-PCMs) when the angle between the resonant rings is α = 105°, while Fig. 3b presents the structure of the linear-to-cross-linear polarization conversion metasurfaces (LTC-LPCMs) with the resonant ring angle set to α = 230°. The polarization conversion metasurface features a sandwich structure with a metal pattern on top and a metallic pattern on the bottom. Each layer has a thickness of 0.035 mm and a conductivity of 5.8 × 10⁷ S/m, effectively blocking wave propagation. The two layers are separated by a dielectric spacer made of FR-4, which has a dielectric constant of 4.3 and a loss tangent of 0.02. To validate the designed metasurface, we employed the Finite Integration Time Domain (FITD) method to model and analyze the metasurface parameters using CST Microwave Studio software. During simulation, an infinite element structure is emulated using a periodic arrangement of cells along the x- and y-axes. Floquet port excitation is set in the z-axis direction to simulate the propagation of EM waves. The size of the optimized metasurface element is defined as follows: period P = 4 mm, thickness d = 1.6 mm, outer radius of top metal resonant ring r = 1.4 mm, ring width w = 0.2 mm, and gap l = 1.7 mm.

Fig. 3
figure 3

(a) The plan view of LTC-PCMs, (b) the plan view of LTCL-PCMs, and (c) side view.

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Polarization conversion parameters

Firstly, simulations were conducted for the cross-polarization reflection coefficient, co-polarization reflection coefficient, and AR of the LTC-PCMs under y-polarized LP wave incidence, as shown in Fig. 4a and b. The results in Fig. 4a indicate that in the frequency bands of 14.6–26.8 GHz and 31–33.5 GHz, the amplitudes of rxy and ryy are nearly equal. The AR is calculated based on the reflection coefficients. The results are shown in Fig. 4b, where the AR is less than 3 dB in frequency bands of 14.6–26.8 GHz and 31–33.5 GHz. The proposed polarization conversion metasurface exhibits excellent linear-circular polarization conversion efficiency, with relative bandwidths of 58.9% and 7.8%.

Fig. 4
figure 4

(a) The co-polarized and cross-polarized reflection coefficients of LTC-PCMs, (b) the corresponding AR, (c) the co-polarized and cross-polarized reflection coefficients of LTCL-PCMs, (d) the corresponding AR.

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Subsequently, simulations were conducted for LTCL-PCMs, resulting in the results shown in Fig. 4c and d. Figure 4c displays that within the frequency range of 13.6–29.8 GHz, the co-polarized reflection coefficient ryy is less than -10 dB, and the cross-polarization coefficient rxy is nearly 0 dB, while successful cross-polarization conversion within this frequency band. Based on the AR results when EM waves are normally incident on the metasurface, as shown in Fig. 4d, the AR exceeds 10 dB within the frequency band of 13.6–29.8 GHz, with a relative bandwidth of 74% for the metasurface. Additionally, at frequencies of 14.18 GHz, 18 GHz, 24.45 GHz, and 29.13 GHz, corresponding to AR values exceeding 35 dB, efficient polarization conversion is achieved, confirming that the waves are fully converted to linearly polarized (LP) waves.

Theoretical analysis

To gain a deeper understanding of the underlying mechanisms, we rotate the xy-coordinate system counterclockwise by 45° to establish a uv-coordinate system for detailed analysis. This involves rotating the top metal of the LTC-PCMs counterclockwise, as shown in Fig. 5a. As the structure rotates, the AR changes accordingly. The results presented in Fig. 5b indicate that when the symmetry axis coincides with the u-axis or v-axis, the AR is less than 3 dB, indicating successful linear-to-circular polarization conversion. However, polarization conversion cannot be achieved when the symmetry axis is not around the u and v axes. Simulation results demonstrate that the conversion performance of the designed metasurface is governed by the u and v axes. Effective polarization conversion can be achieved when the symmetry axis of the metasurface coincides with the u and v axes.

Fig. 5
figure 5

(a) Schematic diagram of metal surface rotation, (b) AR results for different rotation angles.

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We established UV coordinate systems on the LTC PCMs and LTCL PCMs models, as shown in Fig. 6a and c. As an anisotropic structure, the polarization conversion metasurface exhibits a symmetrical u-axis with no cross-polarization reflection component at the incident point. In the reflective metasurface, we assume that the incident wave propagates from the + z direction toward the metasurface along the y direction, with the electric field decomposed into two orthogonal components along the u-axis and v-axis. The relationship between the xy-coordinate system and the uv-coordinate system is represented by a rotation matrix24:

$$\left[ {\begin{array}{*{20}c} x \\ y \\ \end{array} } \right] = Rot\left( {45^{ \circ } } \right)\left[ {\begin{array}{*{20}c} u \\ v \\ \end{array} } \right]$$
(1)
Fig. 6
figure 6

(a) E-field decomposition diagram of LTC-PCMs, (b) co-polarization reflection coefficients and phases of u and v component LTC-PCMs, (c) E-field decomposition diagram of LTCL-PCMs, (d) co-polarization reflection coefficients and phases of u and v component LTCL-PCMs.

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The rotation matrix is represented as:

$$Rot\left( {45^{ \circ } } \right) = \left[ {\begin{array}{*{20}c} {\cos \left( {45^{ \circ } } \right)} & { - \sin \left( {45^{ \circ } } \right)} \\ {\sin \left( {45^{ \circ } } \right)} & {\cos \left( {45^{ \circ } } \right)} \\ \end{array} } \right]$$
(2)

Therefore, we can obtain:

$$\left[ {\begin{array}{*{20}c} u \\ v \\ \end{array} } \right] = Rot^{ - 1} \left( {45^{ \circ } } \right)\left[ {\begin{array}{*{20}c} x \\ y \\ \end{array} } \right]$$
(3)

Therefore

$$Rot^{ - 1} \left( {45^{ \circ } } \right) = \left[ {\begin{array}{*{20}c} {\cos \left( {45^{ \circ } } \right)} & {\sin \left( {45^{ \circ } } \right)} \\ { - \sin \left( {45^{ \circ } } \right)} & {\cos \left( {45^{ \circ } } \right)} \\ \end{array} } \right]$$
(4)

The relationship between the incident E-field and the reflected E-field can be described by the Jones matrix. When the incident electromagnetic wave propagates along the positive z-axis, the Jones matrix for the E-field of the wave incident along the y-axis is represented as:

$$\vec{E}^{i} = \hat{y}\vec{E}^{i} e^{{j\left( {\omega t + k\pi } \right)}} = \vec{E}^{i} \left[ {\begin{array}{*{20}c} 0 \\ {e^{{j\left( {\omega t + kz} \right)}} } \\ \end{array} } \right]$$
(5)

where ω is the angular frequency, k represents the wave number in free space, and \(\vec{E}^{i}\) is the magnitude of the incident wave E-field. Next, we consider the reflection coefficient matrix Ref, where ruu and rvv are the reflection coefficients in the uv-coordinate system. Since there is no cross-polarization conversion for both u-polarized and v-polarized incident waves, we have ruv = rvu = 0. The reflection matrix (Ref) generated by linearly polarized (LP) incident waves can be written as:

$$Ref = \left[ {\begin{array}{*{20}c} {r_{uu} } & 0 \\ 0 & {r_{vv} } \\ \end{array} } \right]$$
(6)

When y-polarized waves are incident on the metasurface, the reflected E-field is represented as:

$$\begin{aligned} \vec{E}^{r} & = Rot^{ - 1} \left( {45^{ \circ } } \right) \cdot Ref \cdot Rot\left( {45^{ \circ } } \right) \cdot \vec{E}^{i} \\ & = \vec{E}^{i} \left[ {\begin{array}{*{20}c} {\cos \left( {45^{ \circ } } \right)} & {\sin \left( {45^{ \circ } } \right)} \\ { - \sin \left( {45^{ \circ } } \right)} & {\cos \left( {45^{ \circ } } \right)} \\ \end{array} } \right]\left[ {\begin{array}{*{20}c} {r_{uu} } & 0 \\ 0 & {r_{vv} } \\ \end{array} } \right]\left[ {\begin{array}{*{20}c} {\cos \left( {45^{ \circ } } \right)} & { - \sin \left( {45^{ \circ } } \right)} \\ {\sin \left( {45^{ \circ } } \right)} & {\cos \left( {45^{ \circ } } \right)} \\ \end{array} } \right]\left[ {\begin{array}{*{20}c} 0 \\ {e^{{j\left( {wt + kp} \right)}} } \\ \end{array} } \right] \\ & = \vec{E}^{i} \left[ {\begin{array}{*{20}c} {\left( {r_{vv} - r_{uu} } \right)\sin \left( {45^{ \circ } } \right)\cos \left( {45^{ \circ } } \right)e^{{j\left( {\omega t + kz} \right)}} } \\ {\left( {r_{uu} \sin^{2} \left( {45^{ \circ } } \right) + r_{vv} \cos^{2} \left( {45^{ \circ } } \right)} \right)e^{{j\left( {\omega t + kz} \right)}} } \\ \end{array} } \right] \\ \end{aligned}$$
(7)

Due to the anisotropy of the metasurface structure material, the co-polarization reflection coefficients ruu and rvv are independent of each other. However, due to low dielectric loss, the magnitudes of ruu and rvv are both close to 1.0, as shown in Fig. 6b and d. In the absence of dielectric losses, define \(\Delta \varphi_{uv} = \varphi_{uu} - \varphi_{vv}\), and establish the following equation:

$$r_{vv} = r_{uu} e^{{ - j\varphi_{uv} }}$$
(8)

Thus, the reflected E-field can be derived as:

$$\vec{E}^{r} = \frac{1}{2}\vec{E}^{i} r_{uu} \left[ {\begin{array}{*{20}c} {e^{{^{{j\left( {\omega t - \Delta \varphi_{uv} + kz} \right)}} }} - 1} \\ {1 + e^{{^{{j\left( {\omega t - \Delta \varphi_{uv} + kz} \right)}} }} } \\ \end{array} } \right]$$
(9)

Using Euler’s formula, it can be deduced that when Δφuv =  ± 90° (and its odd multiples), the components include both sine and cosine terms, resulting in the linear-to-circular polarization conversion; when Δφuv =  ± 180°, the synthesized field forms a straight line along the x-axis, corresponding to the linear polarization state. Thus, a phase difference Δφuv exists between ruu and rvv in the polarization conversion. The phase difference arises from the orthogonal anisotropy of the metasurface unit structure, which excites different intrinsic modes under v- and u-polarized incidence. When a linearly polarized EM wave impinges on the metasurface, both the incident and reflected waves can be decomposed into the u- and v-axis components The u- and v-polarization components of the incident wave are in phase, while the reflected wave may exhibit a significant Δφuv between these components, thereby achieving the desired polarization conversion.

Based on the above conclusions, the reflection coefficients and phase results of the LTC-PCMs and the LTCL-PCMs under u-polarization and v-polarization are shown in Fig. 6b and d. Figure 6b shows that the values of ruu and rvv are close to 1, attributable to dielectric losses, resulting in differences in the co-polarization reflection coefficients. In the frequency bands of 14.6–26.8 GHz and 31–33.5 GHz, the phase difference between φuu and φvv along the u- and v-axes of the LTC-PCMs remains consistently around 90° (or odd multiples of 90°), indicating successful linear-to-circular polarization conversion across this ultra-wide frequency range.

According to Fig. 6d, the phase difference Δφuv between ruu and rvv of the LTCL-PCMs consistently remains around ± 180° in the frequency band of 13.6–29.8 GHz, indicating cross-polarization conversion. At the resonance points of 14.18 GHz, 18 GHz, 24.45 GHz, and 29.13 GHz, where Δφuu =  ± 180° and the corresponding PCR approaches 100%, indicating near-perfect cross-polarization conversion.

Analysis of incident angle stability

Therefore, in practical applications, to ensure that the metasurface exhibits a good electromagnetic response, the angular dependence of the metasurface must be considered. To evaluate the polarization conversion performance of the designed structure under oblique incidence, the AR is used to assess the linear-to-circular polarization conversion at α = 105°, while the PCR is used to evaluate the cross-polarization conversion at α = 230°. The stability of the LTC-PCMs as the incidence angle increases from 0° to 50° is shown in Fig. 7a. Within the 14.6–26.8 GHz frequency range, even at an oblique incidence of 50°, the AR remains below 3 dB in a bandwidth, achieving stable linear-to-circular polarization conversion. However, in the 31–33.5 GHz frequency range, when the incidence angle increases to 30°, the AR changes significantly, resulting in a notable decrease in conversion efficiency. The stability of the incident angle for LTCL-PCMs is depicted in Fig. 7b. With increasing incident angle, the polarization conversion performance of LTCL-PCMs declines gradually. When the incident angle is 50°, a narrow band with PCR greater than 0.9 remains. The simulation results show that the metasurface designed in this paper can still maintain good polarization conversion performance under oblique incidence.

Fig. 7
figure 7

(a) Stability of LTC-PCMs under 50° oblique incidence, (b) stability of LTCL-PCMs under 50° oblique incidence.

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Principles and methods of scattering beam control and RCS reduction

To control the direction of the reflected beams, phase differences are often leveraged to achieve specific beamforming and RCS reduction through the interference and cancellation of waves reflected from adjacent regions. Based on the phase expression of electromagnetic waves:\(E = Ae^{j\phi }\). Where A is the amplitude, ϕ is the phase, and j is the imaginary unit. When the phase difference between two waves is 180°, they will completely cancel each other out because the electromagnetic fields of the two waves will have equal but opposite values at each point, resulting in a total electromagnetic field intensity of zero. Conversely, if the phase difference between two waves is 0 or 360°, the waves will constructively interfere, forming a wave with doubled intensity.

According to the proposed polarization conversion metasurface, two encoding modes, ‘0’ and ‘1’, are defined by rotating the top metal pattern by 90°. It can be observed that the phase of cross-polarized reflection for the ‘0’ encoding units differs from that for the ‘1’ units by ± 180°, as shown in Fig. 8.

Fig. 8
figure 8

(a) The phase maps of two encoding schemes for LTC-PCMs, (b) the phase maps of two encoding schemes for LTCT-PCMs.

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The encoded metasurface is composed of an N*N array of elements, where each array element consists of M*M basic units. When a plane wave is normally incident, the far-field function can be expressed as25:

$$E\left( {\theta ,\varphi } \right) = AF\left( {\theta ,\varphi } \right) \cdot E_{elem} \left( {\theta ,\varphi } \right)$$
(10)

where E(θ, φ) represents the scattered electric field in the far field direction (θ, φ), Eelem(θ, φ) represents the scattering characteristics of each basic unit (i.e., M × M unit). AE(θ, φ) represents the array factor, which can be expressed as:

$$AF\left( {\theta ,\varphi } \right) = \sum\limits_{n = 1}^{N} {\sum\limits_{m = 1}^{M} {e^{{j\left[ {\left( {n - \frac{1}{2}} \right)kd_{x} \sin \theta \cos \varphi + \left( {m - \frac{1}{2}} \right)kd_{y} \sin \theta \sin \varphi + \beta_{nm} } \right]}} } }$$
(11)

βnm represents the phase delay of the m-th basic unit in the n-th array element, where k is the wave number of the incident wave, k = 2π/λ, λ is the wavelength of the incident wave, θ and φ are the polar and azimuthal angles of the observation direction, respectively. This enables specific beamforming and wavefront control.

Based on the far-field scattering principle, different combinations of encoding arrangements are designed, as shown in Fig. 9a–e. Through the design of the array, the incident wave is diffusely reflected. According to the law of conservation of energy, by enhancing the scattering of electromagnetic energy in different directions, the RCS under normal incidence can be effectively reduced. The RCS of the proposed checkerboard surface can be expressed as:

$$\sigma_{c} = \mathop {\lim }\limits_{r \to \infty } 4\pi r^{2} \frac{{\left| {\vec{E}_{x}^{r} } \right|^{2} }}{{\left| {\vec{E}_{x}^{i} } \right|^{2} }}$$
(12)
Fig. 9
figure 9

(a–e) Different encoding unit states, 18 GHz far-field scattering patterns: (f–j) LTC-PCMs, (k–o) LTCL-PCMs, (p–t) LTCL-PCMs with reduced PEC and RCS under different encoding sequences, and (u) LTC-PCMs with reduced PEC and RCS.

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The approximate reduction in RCS of the metasurface designed in this paper compared to a metal plate of the same size is:

$$\sigma \left( {dBsm} \right)_{r} = 10\lg \left[ {\frac{{\mathop {\lim }\limits_{r \to \infty } 4\pi r^{2} {{\left| {\vec{E}_{x}^{r} } \right|^{2} } \mathord{\left/ {\vphantom {{\left| {\vec{E}_{x}^{r} } \right|^{2} } {\left| {\vec{E}_{x}^{i} } \right|^{2} }}} \right. \kern-0pt} {\left| {\vec{E}_{x}^{i} } \right|^{2} }}}}{{\mathop {\lim }\limits_{r \to \infty } 4\pi r^{2} \left| {\vec{E}_{x}^{r} } \right|^{2} }}} \right]$$
(13)

In the equation, σc represents RCS, \(\vec{E}_{x}^{i}\) and \(\vec{E}_{x}^{r}\) denote the incident field and the reflected field in the far field as r approaches infinity, r is the detection distance.

Firstly, we simulate the proposed polarization conversion metasurface in the arrangement shown in Fig. 9a–e. Following the encoding arrangement depicted in Fig. 9a, simulations of the three-dimensional far-field scattering pattern at 18 GHz for LTC-PCMs are conducted, with the results depicted in Fig. 9f. The incident energy is reflected as a primary reflection beam that is opposite in direction to the incident wave, and the number of beams being consistent. Simulated using the bar-coded arrangement depicted in Fig. 9b, the incident energy is reflected as a symmetrical beam. Due to the array’s alignment along the y-axis, the energy of the reflected beam is primarily distributed along the x-axis, as shown in Fig. 9g.

When the metasurface is arranged in a “1001” chessboard coding array, encoding is performed using units of 2*2, 4*4, and 14*14, as illustrated in Fig. 9c–e. In the 2*2 coding configuration, the reflected beam primarily consists of a single main energy beam, which reflects uniformly along the x-axis and y-axis, as depicted in Fig. 9h. When the array is expanded to 7*7 coding units, the reflected beam is evenly divided into five primary energy-level beams, as shown in Fig. 9i, indicating that the increase in coding units leads to a multidirectional dispersion of the reflected energy. When the array is further expanded to 14*14 unit, a noticeable energy convergence occurs in five primary energy-level beams. This is because, compared to the 7*7 unit structure, the 14*14 unit array generates more secondary beams, whose energy gradually accumulates and intensifies, resulting in a more concentrated distribution of the reflected beams, as shown in Fig. 9j. This energy convergence effect highlights the influence of the number of coding units on the distribution of the reflected beams.

Similarly, the far-field simulation is conducted for the cross-polarization metasurface LTCL-PCMs, resulting in different beam patterns. When the encoding is as shown in Fig. 9a–c, both exhibit similar beam effects, as illustrated in Fig. 9k–m. However, significant differences arise when using the “1001” array for encoding. Specifically, under 7*7 coding units, the LTCL-PCMs generate four primary energy-level beams, as shown in Fig. 9n. Upon further expansion to 14*14 coding units, these four high-energy beams exhibit a convergence effect but do not intersect, as depicted in Fig. 9o. This difference is primarily due to the increase in the number of secondary energy-level beams, leading to more concentrated beam energy and a change in distribution characteristics.

Based on looking at the different beams, further investigation was conducted into the phenomenon of reflected wave diffusion in polarization conversion metasurfaces when subjected to linearly polarized beam incidence. By comparing the RCS of metasurfaces and perfect electric conductor (PEC) metal surfaces under different arrangements, the scattering characteristics of the proposed polarization conversion metasurface are verified. For the LTCL-PCMs, in the case of encoding arranged as shown in Fig. 9a, the simulation results indicate in Fig. 9p that the RCS is minimally reduced. When arranged with the encoding in Fig. 9b, there is a reduction in RCS greater than 10 dB in certain frequency bands, corresponding to a change in the beams, as shown in Fig. 9q. However, under the encoding arrangement in Fig. 9c, the small size of the units results in negligible changes in the beam, and thus, the RCS remains largely unchanged, as shown in Fig. 9r. As the array gradually increases, the reduction effect in the 13.6–29.8 GHz band exceeds 10 dB, becoming increasingly significant, as shown in Fig. 9s and t. The reduction in RCS corresponds to the three-dimensional scattering beams shown in Fig. 9n and o, where the incident energy is deflected in various directions. With an increase in the unit structure, the RCS reduction effect becomes even more pronounced.

Similarly, simulations were conducted for LTC-PCMs using the same approach, selecting the encoding arrangement that yielded the best RCS reduction effect from LTCL-PCMs, as shown in Fig. 9e. The simulation results in Fig. 9u indicate that the RCS reduction effect of LTC-PCMs is relatively small, with only some narrow bands exhibiting reductions exceeding 10 dB, which aligns with their far-field beam characteristics. Unlike the four beams of LTCL-PCMs, the beams of LTC-PCMs are not evenly distributed, with a significant energy beam detected at the central point. Although there is a reduction in RCS, the reduction effect is not as excellent as that demonstrated by LTCL-PCMs. Therefore, when designing and applying polarization conversion metasurfaces, it is necessary to consider both their beam characteristics and the reduction effect on RCS to meet specific requirements.

Experiment

To validate the simulation results, we fabricated metasurface samples on a 160 mm × 160 mm dielectric substrate, etched on FR4 using screen printing technology. The internal structure of the metasurface is clearly visible, with Fig. 10a represents the LTC-PCMs sample, Fig. 10b represents the LTCL-PCMs sample, and Fig. 10c represents the LTCL-PCMs sample arranged in a checkerboard pattern. The polarization conversion was experimentally tested in a microwave anechoic chamber, with the experimental setup illustrated in Fig. 10d.

Fig. 10
figure 10

(a) The LTC-PCMs sample, (b) the LTCL-PCMs sample, (c) the LTCL-PCMs sample arranged in a checkerboard pattern, and (d) the experimental testing environment.

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Two horn antennas were connected to a vector network analyzer (N5225A), aligned in both the same and orthogonal directions, respectively, to measure the co-polarized and cross-polarized reflection coefficients. The results for LTC-PCMs and LTCL-PCMs are shown in Fig. 11a and b, demonstrating good agreement between the simulation and measurement results. Then, Fig. 10c employs samples with dimensions of 160 mm × 160 mm × 1.6 mm, fabricated in the arrangement shown in Fig. 9e, to validate the reduction effect on RCS. Figure 11c depicts the simulated and measured RCS values, showing good agreement between the two, both of which are lower than the RCS value of PEC. The experimental results exhibit slight discrepancies compared to the simulation results, attributed to manufacturing tolerances and the experimental environment. Moreover, the simulation employed periodic boundary conditions, while the actual measurement involved finite-sized samples, leading to edge scattering effects.

Fig. 11
figure 11

(a) Simulation and experimental results of reflectance coefficient for LTC-PCMs, (b) simulation and experimental results of reflectance coefficient for LTCL-PCMs, (c) simulation and experimental results of the RCS arranged according to Fig. 9 of LTC-PCMs.

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The scope and functionality of the proposed polarization conversion metasurface in this paper have been compared with previous studies. The specific comparative results are presented in Table 1. The data reveal that the proposed metasurface exhibits excellent performance in terms of ultra-wideband operation and multifunctionality compared to other metasurfaces. Furthermore, a comparison was made between RCS reduction and existing research. Through different reduction methods, the RCS of the metasurface was effectively reduced, demonstrating excellent performance, as shown in Table 2.

Table 1 Compared to the performance of previously proposed polarization conversion metasurfaces.
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Table 2 Compared to the performance of previously proposed RCS reduction metasurfaces.
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Conclusion

In conclusion, this work proposes and fabricates a highly flexible reflective polarization conversion metasurface, which is applied to control beam quantity and reduce RCS. Different conversion effects and bandwidths are achieved at different rotation angles. By rotating through angles of 0 to 2π α, selective frequency band and polarization conversion are achieved within the 12–35 GHz range, enhancing the metasurface’s selectivity and flexibility. Subsequently, by employing various encoding arrays to achieve phase differences in the metasurface, the effectiveness of different scattering beams was verified, successfully leading to RCS reduction. The proposed metasurface demonstrates effective polarization conversion and reliable electromagnetic scattering control across the studied frequency range. Although the design shows promise in enhancing electromagnetic stealth and communication technologies, further research is required to explore its full potential in more complex operational scenarios.